This invention relates to a switching power supply circuit which can be incorporated as a power supply in various electronic apparatus.
A switching power supply circuit which adopts a switching converter in the form of, for example, a flyback converter or a forward converter is widely known. Since switching converters of the types mentioned use a signal of a rectangular waveform as a signal for a switching operation, they have a limitation to suppression of switching noise. It is also known that the switching converters have a limitation to augmentation in power conversion efficiency from their operation characteristics.
Thus, various switching power supply circuits which employ various converters of the resonance type have been proposed by the assignee of the present application. A converter of the resonance type is advantageous in that a high power conversion efficiency can be obtained readily and low noise is realized because the switching operation waveform is a sine waveform. It is advantageous also in that it can be formed from a comparatively small number of parts.
FIG. 10 shows an example of a switching power supply circuit. The switching power supply circuit shown in FIG. 10 includes a rectifier smoothing circuit for rectifying and smoothing the commercial ac power supply AC. The rectifier smoothing circuit is formed as a voltage multiplying rectifier circuit composed of a pair of rectifier diodes Di1 and Di2 and a pair of smoothing capacitors Ci1 and Ci2. The voltage multiplying rectifier circuit produces, for example, where an dc input voltage equal to a peak value of an ac input voltage VAC is represented by Ei, a dc input voltage 2Ei approximately equal to twice the dc input voltage Ei.
The reason why a voltage multiplying rectifier circuit is adopted as a rectifier smoothing circuit in this manner is that it is intended to satisfy the condition of a comparatively heavy load that the ac input voltage is AC 100 V and the maximum load power is 150 W or more.
The switching converter of the voltage resonance type shown in FIG. 10 has a self-excited construction including a single switching element Q1. In this instance, the switching element Q1 may be a high voltage withstanding bipolar transistor (BJT: junction transistor). The base of the switching element Q1 is connected to the positive electrode side of the smoothing capacitor Ci1 (rectified smoothed voltage 2Ei) through a starting resistor RS so that the base current upon starting may be obtained from the rectifier smoothing line. Further, a resonance circuit for self-excited oscillation driving is connected between the base of the switching element Q1 and the primary side ground and is formed from a series connection circuit including an inductor LB, a detection driving winding NB, a resonance capacitor CB, and a base current limiting resistor RB.
A clamp diode DD is interposed between the base of the switching element Q1 and the negative electrode (primary side ground) of the smoothing capacitors Ci and forms a path for damper current which flows when the switching element Q1 is off The collector of the switching element Q1 is connected to an end of a primary winding N1 of an insulating converter transformer PIT, and the emitter of the switching element Q1 is grounded.
A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. The parallel resonance capacitor Cr forms, based on a capacitance of the parallel resonance capacitor Cr itself and a combined inductance (L1+Lc) obtained from a series connection of a leakage inductance L1 of the primary winding N1 side of an orthogonal insulating converter transformer PRT which is hereinafter described and an inductor Lc of a choking coil PCC, a primary side parallel resonance circuit of the voltage resonance type converter. Although detailed description is omitted here, when the switching element Q1 is off, an operation of the voltage resonance type is obtained by an action of the parallel resonance circuit which causes the voltage Vcr across the parallel resonance capacitor Cr to actually exhibit a sine pulse wave.
The choking coil PCC has a transformer coupling construction of the inductor Lc and the detection driving winding NB. The detection driving winding NB excites an alternating voltage corresponding to a switching period in response to a switching output transmitted from the primary winding N1 of the orthogonal insulating converter transformer PRT to the inductor Lc.
The orthogonal insulating converter transformer PRT has a function of transmitting a switching output of the switching element Q1 to the secondary side thereof and performing constant voltage control of the secondary side output thereof The orthogonal insulating converter transformer PRT includes, for example, as shown in FIG. 11, a three dimensional core 200 which is formed such that two double channel-shaped cores 201 and 202 each having four magnetic legs are joined to each other at the ends of the magnetic legs thereof. The primary winding N1 and a secondary winding N2 are wound in the same winding direction around two predetermined ones of the magnetic legs of the three dimensional core 200 and a control winding NC is wound around two predetermined ones of the magnetic legs of the three dimensional core 200 such that the winding direction thereof is orthogonal to the primary winding N1 and the secondary winding N2, whereby the orthogonal insulating converter transformer PRT is formed as a saturable reactor. In this instance, the opposing faces of the opposing legs of the double channel-shaped cores 201 and 202 are joined together and have no gap formed therebetween. Referring back to FIG. 10, one end of the primary winding N1 of the orthogonal insulating converter transformer PRT is connected to the collector of the switching element Q1, and the other end of the primary winding N1 is connected to the positive side of the smoothing capacitors Ci (rectified smoothed voltage 2Ei) through a series connection of the inductor Lc of the choking coil PCC as shown in FIG. 10.
On the secondary side of the orthogonal insulating converter transformer PRT, an alternating voltage induced by the primary winding N1 appears in the secondary winding N2. In this instance, as a secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, a parallel resonance circuit is formed from a leakage inductance L2 of the secondary winding N2 and a capacitance of the secondary side parallel resonance capacitor C2. The alternating voltage induced in the secondary winding N2 is converted into a resonance voltage by the parallel resonance circuit. In short, a voltage resonance operation is obtained on the secondary side.
In the parallel resonance circuit on the secondary side formed in such a manner as described above, center taps are provided for the secondary winding N2, and rectifier diodes D01, D02, D03 and D04 and smoothing capacitors C01 and C02 are connected in such a manner as shown in FIG. 10 to provide two full-wave rectifier circuits including a full-wave rectifier circuit including rectifier diodes D01 and D02 and smoothing capacitor C01 and another full-wave rectifier circuit including rectifier diodes D03 and D04 and smoothing capacitor C02.
The full-wave rectifier circuit composed of the rectifier diodes D01 and D02 and smoothing capacitor C01 receives a resonance voltage supplied from the secondary side parallel resonance circuit and produces a dc output voltage E01. The full-wave rectifier circuit composed of the rectifier diodes D03 and D04 and smoothing capacitor C02 similarly receives the resonance voltage supplied from the secondary side parallel resonance circuit and produces a dc output voltage E02. It is to be noted that, in this instance, the dc output voltage E01 and the dc output voltage E02 are inputted also to a control circuit 1. The control circuit 1 utilizes the dc output voltage E01 as a detection voltage and utilizes the dc output voltage E02 as an operation power supply therefor. The control circuit 1 supplies dc current, whose level thereof varies, for example, in response to the level of the dc output voltage E01 of the secondary side, to the control winding NC of the orthogonal insulating converter transformer PRT to perform constant voltage control in such a manner as hereinafter described.
Since the control winding NC is wound on the orthogonal insulating converter transformer PRT, the orthogonal insulating converter transformer PRT which acts as a saturable reactor operates so that it varies the leakage inductances (L1 and L2). While the leakage inductance L1 of the primary winding N1 forms the parallel resonance circuit of the primary side and the leakage inductance L2 of the secondary winding N2 forms the parallel resonance circuit of the secondary side as described hereinabove, both of the leakage inductances L1 and L2 are variably controlled as the control current flowing through the control winding NC varies as described above. Since the operation just described varies the resonance impedances of the primary side and the secondary side, also the switching output transmitted from the primary side to the secondary side varies, and the secondary side dc voltages (E01 and E02) are controlled to a constant voltage thereby. It is to be noted that such a constant voltage control method as just described is hereinafter referred to as xe2x80x9cparallel resonance frequency control methodxe2x80x9d.
FIG. 12 shows another example of a switching power supply circuit. Also the power supply circuit shown in FIG. 12 employs AC 100 V for a commercial power supply used, for example, in Japan or the United States, similarly to the power supply circuit described hereinabove with reference to FIG. 10, and is ready for the condition that the maximum load power is 150 W or more. Further, a self-excited converter of the voltage resonance type including a single switching element Q1 is provided on the primary side of the switching power supply circuit. It is to be noted that, in FIG. 12, like reference characters to those of FIG. 10 denote like elements and overlapping description thereof is omitted herein to avoid redundancy. Referring to FIG. 12, the switching power supply circuit shown includes an orthogonal control transformer PRT. The orthogonal control transformer PRT includes a three dimensional core 200 formed such that two double channel-shaped cores 201 and 202 each having four magnetic legs are joined to each other at the ends of the magnetic legs thereof. A controlled winding NR is wound by a predetermined number of turns around two predetermined ones of the magnetic legs of the three dimensional core 200. Further, a control winding NC is wound around two predetermined ones of the magnetic legs of the three dimensional core 200 such that the winding direction is orthogonal to the winding direction of the controlled winding NR, whereby the three dimensional core 200 is formed as a saturable reactor.
The orthogonal control transformer PRT can be regarded as a variable inductance element and can be reduced in size when compared with, for example, the orthogonal control transformer PRT described hereinabove with reference to FIG. 11. In this instance, the controlled winding NR is interposed in series between the positive electrode terminal of a smoothing capacitor Ci1 and a primary winding N1 of an insulating converter transformer PIT. Accordingly, in the power supply circuit shown in FIG. 12, a parallel resonance circuit wherein the switching operation of the primary side is of the voltage resonance type is formed by a combined inductance (L1+LR) obtained by a series connection of a leakage inductance L1 of the primary winding N1 side of the insulating converter transformer PIT and an inductance LR of the controlled winding NR and a capacitance of a parallel resonance capacitor Cr.
In the insulating converter transformer PIT shown in FIG. 12, an EE-shaped core 100 is formed from two E-shaped cores 101 and 102 made of a ferrite material, for example, in such a manner as shown in FIG. 14. In this instance, no gap is formed between the central magnetic legs of the E-shaped cores 101 and 102 as seen in FIG. 14. The primary winding N1 (and a detection driving winding NB) and the secondary winding N2 are wound in a separate condition from each other on the central magnetic legs actually using a split bobbin. Further, in the insulating converter transformer PIT, the mutual inductance M between the leakage inductance L1 of the primary winding N1 and the leakage inductance L2 of the secondary winding N2 may have a value +M (additive polarity mode) and another value xe2x88x92M (subtractive polarity mode) depending upon the relationship between the polarities (winding directions) of the primary winding N1 and the secondary winding N2 and the connection to the rectifier diodes D0 (D01 and D02).
The parallel resonance circuit converts the alternating voltage excited in the secondary winding N2 into a resonance voltage. The resonance voltage is supplied to two half-wave rectifier circuits including a half-wave rectifier circuit composed of a rectifier diode D01 and a smoothing capacitor C01 and another half-wave rectifier circuit composed of another rectifier diode D02 and another smoothing capacitor C02. Then, dc output voltages E01 and E02 are obtained from the two half-wave rectifier circuits.
Since the controlled winding NR forms a parallel resonance circuit for obtaining a switching operation of the voltage resonance type as described hereinabove, the resonance condition of the parallel resonance circuit varies with respect to the switching frequency which is fixed. Across the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a resonance pulse of a sine waveform appears by an action of the parallel resonance circuit in response to an off period of the switching element Q1, and the width of the resonance pulse is variably controlled by the variation of the resonance condition of the parallel resonance circuit. In short, a PWM (Pulse Width Modulation) control operation for a resonance pulse is obtained. The PWM control of the resonance pulse width is control of the off period of the switching element Q1, and this signifies, in other words, that the on period of the switching element Q1 is variably controlled in the condition that the switching frequency is fixed. As the on period of the switching element Q1 is variably controlled in this manner, the switching output which is transmitted from the primary winding N1 which forms the parallel resonance circuit to the secondary side varies, and also the output level of the dc output voltage (E01 and E02) of the secondary side varies. Consequently, the secondary side dc output voltage (E01 and E02) is controlled to a constant voltage. It is to be noted that such a constant voltage control method as just described is hereinafter referred to as xe2x80x9cprimary side voltage resonance pulse width control methodxe2x80x9d.
In the switching power supply circuits having the constructions described with reference to FIGS. 10 to 15, in order to satisfy the conditions that the ac input voltage VAC is AC 100 V and the maximum load power is 150 W or more, a dc input voltage of the level of 2Ei is obtained using the voltage multiplying rectification method. Therefore, actually a resonance voltage Vcr of 1,800 V appears between the opposite ends of the switching element Q1 and the parallel resonance capacitor Cr when the switching element Q1 is off. Therefore, for the switching element Q1 and the parallel resonance capacitor Cr, it is required to use products having a withstanding property of the high voltage of 1,800 V. Accordingly, the switching element Q1 and the parallel resonance capacitor Cr have corresponding large sizes. Particularly where a product of a high voltage withstanding property is selected for the switching element Q1, since the saturation voltage VCE(SAT) is high and the storage time tSTG and the fall time tf are long while the current amplification factor hFE is low, it is difficult to set the switching frequency to a high value. As the switching frequency becomes lower, the switching loss and the drive power increase, and consequently, the power loss of the power supply circuit increases. Further, a transformer provided in the power supply circuit and capacitors provided in the driving circuit system increase in size and hence in cost, and this makes an obstacle to reduction in size and weight and reduction in cost of the circuit.
Also in any of the constant voltage control methods described hereinabove with reference to FIGS. 10 and 12, the insulating converter transformer PIT (or the orthogonal insulating converter transformer PRT) wherein the primary side and the secondary side are separate from each other has a required coupling which is obtained without a gap formed therein, and the winding (inductor) Lc of the choking coil or the controlled winding NR of the orthogonal control transformer PRT is connected in series to the primary winding N1 or the secondary winding N2. Consequently, a leakage inductance component in the power supply circuit increases. The increase of the leakage inductance component gives rise to an increase of leakage flux and may possibly have an influence on an electronic circuit and so forth on the load side. Therefore, in order to reduce the influence of leakage flux, actually a structure is adopted wherein, for example, an entire switching converter circuit is accommodated in a shield case made of aluminum and having vent holes formed therein and a connector is provided for connection to inputs and outputs of the switching converter circuit. Also this structure makes an obstacle to reduction in size and weight and reduction in cost of the circuit and increases the time required for manufacture accordingly.
In view of the foregoing, it is thus an object of the present invention to provide an improved switching power supply circuit.
It is a further object of the invention to provide a switching power supply circuit of the resonance type which can achieve promotion of reduction in size and weight and also in cost, augmentation in efficiency in production and augmentation in various characteristics beginning with a power conversion efficiency.
Still other objects and advantages of the invention will in part be obvious and will in part be apparent from the specification and the drawings.
In order to attain the object described above, according to the present invention, there is provided a switching power supply circuit. The switching power supply circuit includes a rectifier smoothing means for receiving a commercial ac power supply, producing a rectified smoothed voltage and outputting the rectified smoothed voltage as a dc input voltage and an insulating converter transformer for transmitting a primary side output to a secondary side where the insulating converter transformer has a gap formed therein so that a coupling which is efficient for a loose coupling is obtained. The switching power supply circuit further includes switching means including a switching element for switching the dc input voltage between on and off states so as to be outputted to a primary winding of the insulating converter transformer, a primary side parallel resonance circuit formed from a leakage inductance component from the primary winding of the insulating converter transformer and a capacitance of a parallel resonance capacitor for enabling the switching means to operate as a voltage resonance type and a secondary side series resonance circuit including a secondary side series resonance capacitor and a secondary winding of the insulating converter transformer, the capacitor connected in series to the secondary winding of the insulating converter transformer, such that a series resonance circuit is formed from a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance of the secondary side series resonance capacitor.
The switching power supply circuit further includes a dc output voltage production means for receiving an alternating voltage obtained at the secondary winding of the insulating converter transformer and performing a voltage multiplying full-wave rectification operation for the alternating voltage to produce a secondary side dc output voltage substantially equal to twice the input voltage level and a constant voltage control means for varying a switching frequency of the switching element in response to a level of the secondary side dc output voltage to perform constant voltage control of the secondary side output voltage.
The switching power supply circuit further includes a series resonance circuit formed from at least a series connection of a driving winding and a resonance capacitor. The switching power supply circuit may further include a self-excited oscillation driving circuit for driving the switching element in a self-excited manner based on a resonance output of the series resonance circuit. The constant voltage control means includes an orthogonal control transformer serving as a saturable reactor on which a detection winding and the driving winding connected in series to the primary winding of the insulating converter transformer, and a control winding whose winding direction is orthogonal to the winding directions of the detection winding and the driving winding are wound, whereby control current which is variable in response to a level of the secondary side dc output voltage is supplied to the control winding to vary the inductance of the driving winding to variably control the switching frequency. The switching means further includes a separately excited driving circuit for driving the switching element in a separately excited manner, and the constant voltage control means variably controls an on period of the switching element in response to a level of the secondary side dc output voltage while keeping an off period of the switching element fixed thereby to variably control the switching frequency.
In the switching power supply circuit, a switching converter of the voltage resonance type is provided on the primary side and the insulating converter transformer is formed such that it has a loose coupling so that operation modes (+M and xe2x88x92M) wherein the mutual inductance between the primary winding and the secondary winding exhibits the opposite polarities to each other may be obtained. Meanwhile, on the secondary side, the secondary side series resonance capacitor is connected in series to the secondary winding to form the series resonance circuit, and the voltage multiplying full-wave rectifier circuit is provided making use of the series resonance circuit so that a secondary side dc output voltage equal to twice an alternating voltage (excited voltage) obtained at the secondary winding may be obtained. Thus, the secondary side dc output voltage is produced by the secondary side series resonance circuit and the voltage multiplying full-wave rectifier circuit to supply power to a load. In short, the voltage multiplying full-wave rectifier circuit is provided basically on the secondary side to cope with a required load condition.
Since power is supplied to the load by the voltage multiplying full-wave rectifier circuit in such a manner as described above, the switching power supply circuit can augment an available maximum load power when compared with, for example, the conventional power supply circuits wherein an equal secondary side dc output voltage is obtained using a full-wave rectifier circuit or a half-wave rectifier circuit. Incidentally, even if, for the primary side, not a voltage multiplying rectifier circuit but an ordinary full-wave rectifier circuit for producing a rectified smoothed voltage equal to the ac input voltage level is provided, the condition described above can be satisfied sufficiently.
Further, in the switching power supply circuit, in order to perform constant voltage control for stabilizing the secondary side output voltage, the switching frequency is varied in response to the secondary side output voltage level to control the resonance impedance of the primary side parallel resonance circuit and the continuity angle of the switching element in the switching power supply circuit simultaneously with each other. Thus, augmentation of the control sensitivity is achieved by the composite control operation.
The above and other objects, features and advantages of the present invention will become apparent from the following description and the appended claims, taken in conjunction with the accompanying drawings in which like parts or elements denoted by like reference symbols.